Soft switched single stage wireless power transfer

ABSTRACT

A control scheme and architecture for a wireless electrical energy transmission circuit employs two solid-state switches and a zero voltage switching (ZVS) topology to power an antenna network. The switches drive the antenna network at its resonant frequency and simultaneously energize a separate resonant circuit that has a resonant frequency lower than the antenna circuit. The resonant circuit creates out of phase voltage and current waveforms that enable the switches to operate with (ZVS).

CROSS-REFERENCES TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.14/588,102 filed Dec. 31, 2014 that claims priority to ProvisionalApplication No. 62/055,191 filed Sep. 25, 2014 titled “SOFT SWITCHEDSINGLE STAGE WIRELESS TRANSFER”, which are hereby incorporated byreference in their entirety for all purposes.

FIELD

The present invention relates generally to wireless power transfercircuits and in particular to the wireless power transfer utilizing anantenna, resonator or resonant tank network.

BACKGROUND

Wireless energy transmission refers to technologies that transmitelectrical energy from a power source to a separate device, such as acell phone or laptop computer without cables or conductors. Twoarchitectures that are widely used for wireless power transfer areantennas (i.e., antenna-based near-field resonator) and inductivecoupling. Antenna-based resonant systems offer better efficiency andgreater power transfer distances than inductive coupling basedtechnologies. The antenna coil typically has a narrow operating range interms of drive voltage, typically 5-40V depending on the power level andcoil design.

Wireless power transmitters typically drive the antenna with a two stagepower converter. The first stage rectifies the AC mains to anintermediate low voltage DC bus (5-40V). The second stage is an inverterthat converts the intermediate low voltage DC bus voltage to AC thatexcites the antenna coil. These power conversion architectures arecomplex, costly and inefficient.

SUMMARY

In one embodiment a wireless power transmission circuit is disclosed.The circuit includes a voltage source having first and second outputterminals that supplies the circuit with power. The circuit furtherincludes a first solid-state switch having a pair of first powerterminals and a first gate terminal. The pair of first power terminalsare connected between the first output terminal of the voltage sourceand a switch node connection. A second solid-state switch has a pair ofsecond power terminals and a second gate terminal. The pair of secondpower terminals are connected between the switch node and the secondoutput terminal of the voltage source.

The circuit further includes an antenna network coupled to the switchnode and configured to transmit electrical energy at a first frequency.A resonant circuit is coupled to the switch node and configured toresonate at a second frequency that is lower than the first frequency. Acontroller is coupled to the first and the second gate terminals and isconfigured to operate the first and the second solid-state switches suchthat they regulate power from the voltage source to drive the antennanetwork at the first frequency. The first frequency interacts with theresonant circuit creating out of phase voltage and current signals atthe switch node. The out of phase signals enable the first and thesecond solid-state switches to operate with zero voltage switching.

In further embodiments the resonant circuit includes a capacitor and aninductor. In yet further embodiments the first and the secondsolid-state switches are GaN-based field-effect transistors. In oneembodiment the voltage source supplies a voltage of 400 volts or greaterat the first and the second output terminals. In some embodiments thefirst frequency is 5 MHz or greater.

In one embodiment the first solid-state switch is turned on by thecontroller after the first pair of power terminals have approximately 0volts across them and in another embodiment the controller will not turnon unless the output capacitance (Coss) of the first solid-state switchis discharged. Some circuits may employ a voltage divider to reduce aninput voltage supplied to the antenna network. In further embodimentsthe antenna network is configured to resonate at the first frequency.

In some embodiments the circuit may further include a third solid-stateswitch having a pair of third power terminals and a third gate terminal.The pair of third power terminals are connected between the first outputterminal and a second switch node. The circuit may further have a fourthsolid-state switch having a pair of fourth power terminals and a fourthgate terminal. The pair of fourth power terminals are connected betweenthe second switch node and the second output terminal. The antennanetwork and the resonant circuit are connected between the switch nodeand the second switch node. The controller is coupled to the third andthe fourth gate terminals such that it can drive all four switches.

In yet further embodiments the circuit may further comprise a secondresonant circuit disposed between the switch node and the second switchnode, connected in parallel with the resonant circuit. In someembodiments the circuit may include a second resonant circuit coupled inparallel with the second switch and connected between the switch nodeand the second output terminal; and a third resonant circuit coupled inparallel with the fourth switch and connected between the second switchnode and the second output terminal.

Further embodiments include a method of operating a wireless powertransmission circuit. The method includes supplying power to the circuitwith a voltage source having a first and a second output terminal. Afirst gate control signal is transmitted to a first driver circuit. Inresponse, the first driver circuit transmits a first gate drive signalto a gate of a first solid-state switch. The first solid-state switchhas a pair of first power terminals connected between the first outputterminal and a switch node. A second gate control signal is transmittedto a second driver circuit. In response, the second driver circuittransmits a second gate drive signal to a gate of a second solid-stateswitch.

The second solid-state switch has a pair of second power terminalsconnected between the second output terminal and the switch node. Acontroller transmits the first and the second gate control signals suchthat the first and second solid-state switches turn on and off at afirst frequency, regulating power delivered to the switch node. Anantenna network is coupled to the switch node and driven at the firstfrequency such that electrical energy is radiated from the antenna. Aresonant circuit is coupled to the switch node and energized with thefirst frequency such that out of phase voltage and current signals arecreated. The out of phase signals enable the first and the secondsolid-state switches to operate with zero voltage switching.

In some embodiments a wireless power receiver circuit includes arectifier circuit having first and second input terminals and first andsecond output terminals. The rectifier circuit further includes a firstleg connected between the first input terminal and the first outputterminal, a second leg connected between the first output terminal andthe second input terminal, a third leg connected between the first inputterminal and the second output terminal, a fourth leg connected betweenthe second output terminal and the second input terminal, and at leastone switch disposed within at least one of the first, the second, thethird and the fourth legs. The receiver circuit also includes a receivercoil having a first terminal connected to the first input terminal and asecond terminal connected to the second input terminal. A controller iscoupled to the at least one switch and is configured to control outputpower of the wireless power receiver circuit.

In further embodiments two or more switches may be disposed within atleast one of the first, the second, the third and the fourth legs andcontrolled with the controller. In yet further embodiments, one of theat least one switches may be GaN-based.

In some embodiments a method of operating a wireless power receivercircuit may include receiving AC power with a receiver coil. Thereceiver coil has first and second receiver terminals connected to firstand second input terminals, respectively, of a rectifier circuit. Therectifier circuit may be used to rectify the received AC power and mayinclude a first leg connected between the first input terminal and afirst output terminal, a second leg connected between the first outputterminal and the second input terminal, a third leg connected betweenthe first input terminal and a second output terminal, a fourth legconnected between the second output terminal and the second inputterminal, and at least one switch disposed within at least one of thefirst, the second, the third and the fourth legs. The rectifier circuitis used to regulate DC power at the first and second output terminals byoperating the at least one switch with a controller.

In further embodiments two or more switches may be disposed within atleast one of the first, the second, the third and the fourth legs andcontrolled with the controller. In yet further embodiments, one of theat least one switches may be GaN-based.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic of a simplified single-stage antenna drive networkcircuit in accordance with an embodiment of the invention;

FIG. 2 is schematic of a simplified single-stage antenna drive networkcircuit including a controller and FET drivers in accordance with anembodiment of the invention;

FIG. 3A is schematic of a simplified single-stage antenna drive networkcircuit in accordance with an embodiment of the invention;

FIG. 3B is an example voltage plot of the gate drive voltage of the highside FET in the circuit illustrated FIG. 3A according to an embodimentof the invention;

FIG. 3C is an example voltage plot of the gate drive voltage of the lowside FET in the circuit illustrated in FIG. 3A according to anembodiment of the invention;

FIG. 3D is an example voltage plot of the switch-node voltage of thecircuit illustrated in FIG. 3A according to an embodiment of theinvention;

FIG. 3E is an example current plot of the inductor current in thecircuit illustrated in FIG. 3A according to an embodiment of theinvention;

FIG. 4 is a schematic of a simplified single-stage antenna drive circuitwith a resonant circuit according to an embodiment of the invention;

FIG. 5 is a schematic of a simplified single-stage antenna drive circuitwith a resonant circuit and a voltage divider according to an embodimentof the invention;

FIG. 6 is a schematic of a simplified single-stage antenna drive circuitwith a resonant circuit and a voltage divider according to an embodimentof the invention;

FIG. 7 is a schematic of a simplified single-stage antenna drive circuitwith a resonant circuit and a voltage divider according to an embodimentof the invention;

FIG. 8 is a schematic of a simplified full-bridge single-stage antennadrive circuit with a resonant circuit according to an embodiment of theinvention;

FIGS. 9A-9C are schematics of various antenna network configurationsaccording to embodiments of the invention;

FIG. 10 is a schematic of a simplified single-stage antenna drivenetwork circuit that uses an AC current source in accordance with anembodiment of the invention;

FIG. 11 is a schematic of a simplified single-stage antenna drivenetwork circuit including a pre-regulator according to an embodiment ofthe invention;

FIG. 12 is a schematic of a simplified single-stage antenna drivenetwork circuit that uses a duty cycle algorithm to regulate the averagewireless transmitter output power according to an embodiment of theinvention;

FIG. 13 is an example of a duty cycle algorithm that may be employed forthe circuit described in FIG. 12 according to an embodiment of theinvention;

FIG. 14A is an example of a phase shift duty cycle algorithm that may beemployed for the circuit described in FIG. 12 according to an embodimentof the invention;

FIG. 14B is a schematic of a simplified single-stage antenna drivenetwork circuit that uses a duty cycle algorithm and additional resonantcircuits to achieve ZVS and regulate the average wireless transmitteroutput power according to an embodiment of the invention;

FIG. 14C is a schematic of a simplified single-stage antenna drivenetwork circuit that uses a duty cycle algorithm and additional resonantcircuits to achieve ZVS and regulate the average wireless transmitteroutput power according to an embodiment of the invention;

FIG. 15 is an example of an antenna coil according to an embodiment ofthe invention;

FIG. 16 is a schematic of a single-stage antenna drive circuit with aresonant circuit according to an embodiment of the invention;

FIG. 17 is a schematic of a single-stage antenna drive circuit with aresonant circuit according to an embodiment of the invention;

FIG. 18 is a schematic of a receiver network according to an embodimentof the invention;

FIG. 19 is a schematic of the receiver network illustrated in FIG. 18including a DC to DC converter according to an embodiment of theinvention;

FIG. 20 is a schematic of receiver network with synchronous switchesused in place of diodes for a bridge rectifier according to anembodiment of the invention;

FIGS. 21A-21B are schematics of bidirectional switches according toembodiments of the invention;

FIG. 22 is a schematic of receiver network having two bidirectionalswitches combined with a synchronous rectifier to regulate DC outputaccording to an embodiment of the invention;

FIG. 23 is a schematic of receiver network having two bidirectionalswitches combined with a synchronous rectifier to regulate DC outputaccording to an embodiment of the invention;

FIG. 24 is a schematic of receiver network having two bidirectionalswitches combined with a synchronous rectifier to regulate DC outputaccording to an embodiment of the invention;

FIG. 25 is a schematic of receiver network having a single stageregulation architecture according to an embodiment of the invention;

FIG. 26 is a simplified isometric rendering of co-packaged devicesaccording to an embodiment of the invention; and

FIG. 27 is a simplified isometric rendering of co-packaged devicesaccording to an embodiment of the invention.

DETAILED DESCRIPTION

Certain embodiments of the present invention relate to antenna drivenetworks for wireless electrical energy transmitters. While the presentinvention can be useful for a wide variety of antenna drive networks,some embodiments of the invention are particularly useful for antennadrive networks that use single-stage power conversion and/or zerovoltage switching (ZVS), as described in more detail below.

Many electronic devices such as smart-phones, media players, and tabletcomputers are rechargeable and require charging of a battery to operatewithout a power cord. Some electronic devices may be configured to becharged with a wireless electrical energy transfer system. Wirelesselectrical energy transfer systems typically consist of a powertransmitter that transmits electrical energy through the air to areceiver located within the electronic device. The power transmitter mayreceive power from AC mains to power an antenna drive network that inturn powers the transmitter's antenna. The antenna wirelessly transmitselectrical energy to a receiver within the device. The receiver thenconverts the energy to a usable form and supplies it to the device.

Now referring to FIG. 1, a simplified block diagram of an embodiment ofa single stage antenna drive system 100 is illustrated. A power source105 may be used to supply power to a power regulator 110. Powerregulator 110 may comprise one or more solid-state switches, among otheractive and passive devices, and be configured to drive antenna network115 at a first frequency. The first frequency may interact with resonantcircuit 120 creating out of phase voltage and current signals at powerregulator 110. The out of phase voltage and current signals may be usedto enable ZVS of the one or more solid-state switches in power regulator110, as discussed in more detail below. Controller 125 may control theoperation of the one or more solid-state switches in power regulator 110through driver circuit 130.

Now referring to FIG. 2, one embodiment of a simplified single-stageantenna drive network 200 is illustrated. This embodiment is powered byan AC mains source 205 that is connected to full-wave rectifier 210.Rectifier 210 converts AC from AC mains source 205 to a DC source havingcyclical voltage variations at approximately twice the frequency of theAC mains source. A smoothing capacitor 230 may be used to smooth thecyclical voltage variations, creating a relatively steady high voltageDC power source. DC power is delivered to half-bridge power regulator215 through first output terminal 210 a and second output terminal 210b.

Power regulator 215 comprises first switch 220, second switch 225 andsmoothing capacitor 230. First switch 220 has a pair of first powerterminals 220 a, 220 b connected between the first output terminal 210 aand a switch node 255. Second switch 225 has a pair of second powerterminals 225 a, 225 b connected between the switch node 255 and secondoutput terminal 210 b. Power regulator 215 is controlled by controller235 that operates first and second switches 220, 225, respectively,through first driver circuit 240 and second driver circuit 245.

First driver circuit 240 is connected to first gate 220 c of firstswitch 220. Second driver circuit 245 is connected to second gate 225 cof second switch 225. Controller 235 operates first and second switches220, 225, respectively, at a first frequency that may be called theswitching frequency or (fsw). Resonant circuit 250 is coupled to powerregulator 215 and has a resonant frequency (f0) at a second frequencythat is lower than the first frequency (fsw). In some embodiments,resonant circuit 250 may include an inductor 260 and a capacitor 265,however other embodiments may comprise different components. Theinteraction of the first frequency (fsw) with resonant circuit 250 maycreate out of phase voltage and current signals at switch node 255 thatenable first and second switches, 220, 225, respectively, to operateusing ZVS, as described in more detail below.

Antenna network 267 is coupled to switch node 255 of power regulator 215through resonant circuit 250. Thus, first and second switches 220, 225,respectively, may be switched to regulate power delivered to antennanetwork 267 and/or to drive the antenna network with the appropriate ACfrequency, which may be the antenna network's resonant frequency. Asillustrated, single-stage antenna drive network 200 may power antennanetwork 267 without using an intermediate low voltage DC bus.

As used herein, ZVS means that the semiconductor switch may be turned onor off only when the voltage applied across the switch is at or nearzero (i.e., zero voltage switching or ZVS) and when the outputcapacitance, or Coss, is at or near zero charge. Switching losses (i.e.,turning a switch off while it is conducting current or turning a switchon when it has a voltage potential across it) may be a significantcontributor to power loss in the system. The use of ZVS may result inreduced switching losses, increased frequency of operation and in someembodiments, reduced electromagnetic interference (EMI) generation.

Now referring to FIG. 3A, an example half bridge circuit 300 isillustrated along with selected voltage and current plots (FIGS. 3B-3E)to show how a resonant circuit may create out of phase voltage andcurrent signals enabling ZVS. Circuit 300 has a DC voltage source 305 at300 VDC. First switch 220 is also known as a “High Side” switch and iscontrolled by a gate signal Vgs_H. Second switch 225 is also known as a“Low Side” switch and is controlled by a gate signal Vgs_L. First andsecond switches 220, 225, respectively, are coupled to switch node 310and control power delivered to resonant circuit 315 and a loadrepresented by resistor 320. In some embodiments resistor 320 may be anantenna network.

In this embodiment the switching frequency of first and second switches220, 225, respectively is above a resonant frequency of resonant circuit315. In this example, the resonant frequency of resonant circuit 315 maybe 2 MHz and first and second switches 220, 225, respectively, mayoperate at a switching frequency of approximately 5 MHz. Otherembodiments may operate at different resonant and switching frequencies.

Now referring simultaneously to FIGS. 3A-3E, selected voltage andcurrent waveforms of circuit 300 are plotted during a representativeswitching cycle. As illustrated in FIGS. 3D and 3E, current in inductor360 lags the voltage at switch node 310. That is, the current ininductor 360 (see FIG. 3E) reaches peak current later than the voltageat switch node 310 (see FIG. 3D) reaches peak voltage. Thus circuit 300may be called “inductive”. In embodiments that employ ZVS, the laggingcurrent in an “inductive” circuit may be used to provide energy todischarge output capacitance (Coss) of the first and second switches220, 225, respectively before they turn on, as discussed in more detailbelow.

FIG. 3C plots the gate voltage of second switch 225. At time t1, secondswitch 225 turns off (i.e., its gate voltage transitions from 6 volts to0 volts). As illustrated in FIG. 3B, first switch 220 is also off attime t1. This state causes a negative current in inductor 360 thatdischarges the output capacitor (Coss) in first switch 220. Asillustrated in FIG. 3D, after time t1 the voltage at switch node 310rises quickly to the voltage of source 305 (i.e., 300 volts). Onceswitch node 310 is at the voltage of source 305 there is zero voltagepotential across first switch 220 since both terminals of the firstswitch are at Vin, (i.e., 300V). Thus, ZVS of first switch 220 may beachieved since it now has 0 volts across it and its output capacitance(Coss) has been discharged.

Now referring to FIG. 3B, at time t2, first switch 220 turns on with ZVS(i.e., its gate voltage transitions from 0 volts to 6 volts).Simultaneously at time t2, FIG. 3E shows current in inductor 360increasing towards a peak current. Now referring to FIG. 3B, at time t3first switch 220 is turned off (i.e, its gate voltage transitions from 6volts to 0 volts). Positive current in inductor 360 discharges theoutput capacitance (Coss) of second switch 225. Simultaneously, at timet3, FIG. 3D shows the voltage at switch node 320 decreasing to 0 volts.When switch node 320 voltage reaches 0 volts, the voltage across secondswitch 225 is zero. Thus, ZVS of second switch 225 may be achieved sinceit now has 0 volts across it and its output capacitance (Coss) has beendischarged. Now referring to FIG. 3C, at time t4, second switch 225turns on with ZVS (i.e., its gate voltage transitions from 0 volts to 6volts).

In some embodiments that use ZVS the resonant frequency of resonantcircuit 315 may be between 10% and 60% lower than the switchingfrequency. In such embodiments, first and second switches 220, 225,respectively, may inject a square voltage waveform into resonant circuit315, at node 310. A square voltage waveform at the switching frequency(fsw) may have many harmonic frequencies such as 2fw, 3fsw, 4fsw, etc.that cause the antenna to transmit related harmonic frequencies. In someembodiments, it may not be desirable for the antenna to transmitharmonic frequencies due to communication regulations that require atransmitter to transmit in a narrowly defined bandwidth. Thus, in someembodiments it may be desirable to pass the switching frequency (fsw) tothe antenna with minimal harmonics.

When operating at a switching frequency closer to the resonant frequencyof 315, the voltage and current waveforms going through capacitor 365and inductor 360 may be more sinusoidal, as the network has a highquality factor that serves as a filter. In these embodiments, the squarewave waveform may be filtered through 315 and the voltage waveformapplied to antenna network 267 (see FIG. 2) winding is also moresinusoidal. This may help reduce the injection of harmonic frequenciesinto antenna network 267 (see FIG. 2), improving the performance of theantenna network and assist in meeting regulatory requirements such aselectromagnetic compliance standards.

In further embodiments, the resonant frequency of resonant circuit 315may be much lower than the switching frequency. For example, in someembodiments the resonant frequency of resonant circuit 315 may bebetween 50% to 90% lower than the switching frequency. In theseembodiments, resonant circuit 315 may resemble a low pass filter withoutany, or with little resonant action between Lr and Cr, and ZVS may beachieved for a wide range of load conditions. In these embodiments, thewaveform applied to the antenna network may have a more triangularcurrent shape (see FIG. 3E). Since this waveform is not a puresinusoidal waveform, it too will create harmonics that will betransmitted by the antenna. In such embodiments an additional filter maybe employed in front of antenna network 267 (see FIG. 2), to remove theharmonic content.

Other embodiments may employ ZCS switching such that each switch mayturn off when its drain current reduces to zero. In embodiments thatemploy ZCS circuit architectures, the current leads the voltage signal,which may also be called “being capacitive” and may be achieved by usingLr and Cr values that resonate at a higher frequency than the switchingfrequency. ZCS reduces turn-off switching loss, which may be asignificant energy loss component at high frequency or at high voltage.A similar series resonant circuit (i.e., such as resonant circuit 315)may be used achieve ZCS. In some embodiments, certain types of switches,such as, for example IGBTs may be particularly, useful for ZCS switchesdue to their long current tail during switching transitions.

In further embodiments it may be beneficial for the switches to be ableto withstand high voltage potentials and to switch at high frequencies,particularly when the transmitter runs off AC mains. In one embodimentthe voltage potential across the switches may be in the range of 50-1000VDC and in another embodiment in the range of 100-400 VDC. In otherembodiments the switching frequency may be in the range of 1-20 MHzwhile yet further embodiments may operate in a range between 5-14 MHz.In one embodiment the switching frequency may be 6.78 MHz.

In some embodiments, one or more of the switches may be a silicon-basedMOSFET device. In further embodiments one or more of the switches may bea GaN-based device. In one such embodiment one or more of the devicesmay be fabricated on a substrate having a base of silicon with anepitaxially deposited layer of GaN. In other embodiments differentsubstrate configurations may be employed.

As discussed in more detail below, GaN-based switches may beparticularly useful in embodiments that may be used to efficientlyswitch high voltage buses (e.g., 400V and above) and/or at highfrequencies (e.g., 5-14 MHz). However, none of the embodiments hereinshall be limited in scope to any particular type of switch. Allembodiments may use silicon-based, GaN-based, or a combination thereoffor any solid-state switch. In some embodiments the efficiency of theantenna drive network may be in the range of 70% to 95%. In oneembodiment the efficiency of the antenna drive network may beapproximately 90%.

In some embodiments, GaN based switches may be particularly well suitedto switch at high frequencies due to their lower output capacitance, orCoss values, their relatively fast switching speed and their lower gatecharge requirements. Each time the FET turns on, the energy stored inthe output capacitance will be dissipated in the device. As theswitching frequency increases, the power dissipation in the FET due todischarging this energy increases proportionately, which may become alimiting factor in hard switching topologies. Thus, because GaN-basesswitches have a lower Coss value than silicon, they can operate athigher frequencies with less power dissipation.

Further, silicon-based devices have a body diode that is made from a P-Njunction. When the diode conducts it needs time for charge recovery orit won't be able to block a reverse voltage. As a general rule, thehigher the voltage the longer the recovery time for the body diode. Forexample, at frequencies in the 100 kHz range the diode needs to recoverwithin about 1 microsecond. However, for example, at frequencies in the6 MHz range, the diode needs to recover within approximately 30nanoseconds, but it can't. Therefore, because GaN switches do not havean internal body diode that needs to recover, they can switch faster.Moreover, since GaN-based switches have lower gate charge and lower Cossthan silicon-based switches, it is easier to recover energy since thereis less energy to recover and it takes less time to recover it.

As a non-limiting example, in some embodiments the relatively small Cossassociated with GaN devices, on the order of 2 picofarads, may enablefaster discharging of the Coss and thus higher switching frequencies. Asa further non-limiting example, in some embodiments GaN-based switchesmay switch within approximately 2 nanoseconds, enabling them to operateat high frequencies.

Further, GaN-based devices may be operated with relatively small drivercircuits, even at high voltages, because of their relatively small gatecharge requirements. The smaller charge requirements may make thereduced size and cost of the driver circuit attractive for high voltageapplications. Yet further, since GaN switches have a lateralconstruction, and the driver circuit may be relatively small, someembodiments may benefit from integrating the driver circuitmonolithically or co-packaged with one or more of the FETs, as discussedin more detail below.

Now referring to FIG. 4 an embodiment of a simplified single-stageantenna drive network 400 attached to an antenna network 405 isillustrated. The circuit in FIG. 4 is similar to the circuit in FIG. 2,however antenna network 405 has been added, along with a capacitor 410which is employed as a voltage divider, as discussed in more detailbelow.

In some embodiments antenna network 405 may have its own inductor 420and capacitor 420 forming a resonant antenna circuit. In furtherembodiments, power regulator 215 may operate at a switching frequencymatched to the antenna network 405 resonant frequency such that theantenna network transmits energy for wireless charging. Power regulator215 may simultaneously energize resonant circuit 250 such that itresonates at a different frequency from the switching frequency creatingout of phase voltage and current signals that enable first and secondswitches, 220, 225, respectively, to operate using ZVS. As discussedabove, in one embodiment, resonant circuit 250 may be designed tooperate at a frequency below the switching frequency and first andsecond switches, 220, 225, respectively, operate with ZVS. In anotherembodiment, resonant circuit 250 may be designed to operate at afrequency above the switching frequency and first and second switches,220, 225, respectively, operate with ZCS.

In further embodiments a voltage divider circuit may be used to adjustthe voltage applied to antenna network 405. In FIG. 4 a capacitivevoltage divider circuit is used were the ratios between capacitor 265and capacitor 410 determine the voltage applied to antenna network 405.More specifically capacitor 410 may be used to reduce the AC voltagefrom capacitor 265 to match the requirements of antenna network 405. Insome embodiments capacitor 410 may be larger than capacitor 265. Theratio between capacitors 265 and 410 can be selected such that theresulting voltage across capacitor 410 matches the voltage requirementfor antenna network 405. In one embodiment capacitor 410 is selectedthat is five times bigger than capacitor 265 and the AC voltage suppliedto antenna network 405 is reduced from +/−100V to +/−20V. In otherembodiments a different type of voltage divider circuit or technique toreduce the voltage applied to antenna network 405 may be used withoutdeparting from the invention.

Now referring to FIG. 5, circuit 500 is similar to circuit 400illustrated in FIG. 4, however instead of using a capacitive voltagedivider circuit, circuit 500 uses an inductive voltage divider circuit.More specifically, the ratio between inductor 260 and inductor 505determines the voltage applied to antenna network 405. Thus, inductor505 may be used to reduce the AC voltage from inductor 260 to match therequirements of antenna network 405. For example, if inductor 505 is tentimes smaller than inductor 260, the voltage across capacitor 505 wouldbe approximately ten times smaller than voltage across inductor 260.Myriad methods may be used to adjust the voltage to meet therequirements of antenna network 405 without departing from theinvention.

Now referring to FIG. 6, circuit 600 is similar to circuit 500illustrated in FIG. 5, however, where capacitor 265 (in FIG. 5) isblocking the DC voltage of the input bus, circuit 600 in FIG. 6 uses twocapacitors (605, 610) to divide the input voltage so that antennanetwork 405 is supplied with half the input voltage. In someembodiments, capacitors 605, 610 may make it easier for a start-upcircuit to power antenna network 405. Some embodiments withoutcapacitors 605, 610 may have to turn on the PWM duty cycle gradually toproperly power antenna network 405. The remainder of circuit 600 is thesame as discussed with regard to circuit 500 illustrated in FIG. 5,including using inductors 260 and 505 to reduce the voltage applied toantenna network 405.

Now referring to FIG. 7, circuit 700 is similar to circuit 600illustrated in FIG. 6, however, instead of using an inductive voltagedivider, circuit 700 uses two capacitors (265, 705) to adjust the inputvoltage to antenna network 405.

Now referring to FIG. 8, circuit 800 illustrates a full-bridge circuitthat may be used in some embodiments to convert a high voltage to a lowvoltage for antenna network 405. Full-bridge circuit 800 may beconsidered an extension of the half-bridge circuits discussed in theembodiments above (FIGS, 2, 3A, 4-7), however a third switch 805 and afourth switch 810 may extend the power level such that the full busvoltage is applied to resonant circuit 250. More specifically, thehalf-bridge embodiments described above (FIGS, 2, 3A, 4-7) may applyonly 50 percent of the bus voltage to the resonant circuit whereas fullbridge 800 may apply 100 percent of the bus voltage. In some embodimentsthe lower the bus voltage that is supplied to resonant circuit, the moredifficult the duty cycle is to control, which may be alleviated byadding a pre-regulator. Similar to the embodiments described above, inone embodiment, inductor 260 and capacitor 265 of resonant circuit 250may be tuned to achieve ZVS for first, second, third and fourthswitches, 220, 225, 805, 810, respectively. In further embodimentsantenna network 405 may be tuned to the switching frequency of thefull-bridge circuit to maximize power transfer. In some embodiments, asdiscussed above, matching components such as inductor 505 in FIG. 6 orcapacitor 705 in FIG. 7 may be used to adjust the match of the circuitto antenna network 405.

Now referring to FIGS. 9A-9C, various antenna network configurations(905, 910, 915) are illustrated. Any of these antenna networks may beused in place of antenna network 405 discussed in FIGS. 1-8 above, or inother embodiments. Antenna network 905 in FIG. 9A has a capacitor 906and an inductor 907 in series. Network 905 may be particularly useful inembodiments that have an AC voltage source applied to the antennanetwork. Antenna network 910 in FIG. 9B has a capacitor 911 and aninductor 912 connected in parallel. Network 910 may be particularlyuseful when an AC current source is fed into the antenna network.Antenna network 915 in FIG. 9C may have a capacitor 916 connected inseries with a transformer 917. Network 915 may use transformer 917 totranslate AC voltage to match the antenna network impedance. Myriadantenna networks are within the scope of this disclosure and may be usedwithout departing from the invention.

Now referring to FIG. 10, circuit 1000 illustrates a resonanthalf-bridge circuit that may be used in some embodiments to convert ahigh voltage to a low voltage for an antenna network 405. Circuit 1000is similar to those described above, however instead of feeding ACvoltage to antenna network 405, AC current is fed into the antennanetwork. In some embodiments, resonant circuit 250 is tuned to such thatfirst and second switches 220, 225, respectively are operated using ZVS.

Now referring to circuit 1100 in FIG. 11, an embodiment including apre-regulator 1105 to remove AC ripple and/or to modulate antennanetwork 405 output power is illustrated. As compared to the embodimentsabove that all use single-stage power conversion, this embodimentemploys pre-regulator 1105 which may add flexibility in regulating theamount of power transfer. In one embodiment, pre-regulator 1105 mayinclude a pre-regulator switch 1110, a diode 1115 and an inductor 1120.Other embodiments may have different components and configurations. Insome embodiments pre-regulation may be achieved by modulating the DCvoltage for the inverter circuit. In further embodiments, DC/DCpre-regulator 1105 may be used to step down the voltage to remove linefrequency ripples and modulate the DC voltage fed into the antennadriver to control the wireless power transfer. Such embodiments maydecrease efficiency and increase the cost of the power regulator.

Now referring circuit 1200 in FIG. 12, another embodiment may employ acircuit architecture having a duty cycle algorithm to regulate theaverage wireless transmitter output power from antenna network 405. Inthis embodiment, third and fourth switches 1205, 1210, respectively maybe added and used in conjunction with first and second switches, 220,225, respectively, as discussed in more detail below.

Now referring to FIG. 13, in one embodiment a duty cycle algorithm maybe employed for circuit 1200 described in FIG. 12 that uses dead timebetween first switch pair including first switch 220 and fourth switch1210, and second switch pair including second switch 225 and thirdswitch 1205 to modulate the average power delivered to antenna network405. Waveform 1320 shows the gate voltage of first switch 220; waveform1310 shows the gate voltage of fourth switch 1210; waveform 1325 showsthe gate voltage of second switch 225; waveform 1305 shows the gatevoltage of third switch 1205; and waveform 1330 shows the effectivevoltage applied to antenna network 405 (see FIG. 12).

Continuing to refer to FIG. 13, it can be seen that first switch 220 andfourth switch 1210 turn on and off synchronously. It can also be seenthat second switch 225 and third switch 1205 turn on and offsynchronously. Now referring to circuit 1200 in FIG. 12, it can be seenthat when first switch 220 and fourth switch 1210 turn on, current mayfrom full-bridge rectifier 1215, through first switch 220, throughresonant circuit 250 and antenna network 405 circuit elements, andthrough fourth switch 1210. Conversely, when second switch 225 and thirdswitch 1205 are on, a reverse voltage potential is created and currentmay from full-bridge rectifier 1215, through second switch 225, throughresonant circuit 250 and antenna network 405 circuit elements, andthrough third switch 1205. By regulating the “on time” of the firstswitch pair and the second switch pair, the average power delivered toantenna network 405 can be controlled. The effective voltage of thisoperation is shown by line 1330 in FIG. 13.

Continuing to refer to FIG. 13, it can also be seen that the dead time,also called the off time of the first and the second switch pairs iswhen the opposite switch pair is turned on. For example, at the leftmost position in FIG. 13, second switch 225 and third switch 1205 areon, while first switch 220 and fourth switch 1210 are off. Progressingto the right, second switch 225 and third switch 1205 are off for timet1. During time t1, first switch pair including first switch 220 andfourth switch 1210 are turned on for time t2, then turned off.Subsequently, second switch pair, including second switch 225 and thirdswitch 1205, is turned on, during the time first switch pair includingfirst switch 220 and fourth switch 1210 are off. Thus, only one switchpair is on at a time, and the time the switch pair is on is the amountof time that the transmitter circuit has power applied to it, thusregulating the average power supplied to the circuit. By changing theduty cycle of each switch, the average AC voltage applied to theresonant network is modulated.

Now referring to FIG. 14A, another embodiment employing a phase shiftalgorithm may be used to regulate the average power delivered totransmitter network 405 (see FIG. 12) is illustrated. Similar to thealgorithm applied in FIG. 13, voltage will be applied to antenna network405 only when first switch pair including first switch 220 and fourthswitch 1210 are on or when second switch pair including second switch225 and third switch 1205 are on are. Therefore, when first switch 220and third switch 1205 are on, no power is transferred. Similarly, whensecond switch 225 and fourth switch 1210 are on, no power istransferred.

Further, in some embodiments, first switch 220 and second switch 225 mayswitch complementary, as may third switch 1205 and fourth switch 1210.More specifically, in some embodiments when first switch 220 is on,second switch 225 is off and similarly when third switch 1205 is onfourth switch 1210 may be off. Therefore, in one embodiment all switchesmay operate at a fixed 50% duty cycle (i.e., on for half the time andoff for half the time). Instead of modulating the effective AC voltageapplied to the resonant circuit by changing the duty cycle of eachswitch as discussed above, the AC voltage is modulated by changing thephase delay between the switches. Maximum overlap (zero phase delay)produces a maximum duty cycle and maximum power to antenna network 405.In one embodiment illustrated in FIG. 14A the duty cycle of the powerapplied to antenna network 405 may be controlled by modulating the phasedelay between first switch 220 and third switch 1205.

In some embodiments a phase shifted control methodology may be easierthan other control methodologies to maintain ZVS over large load and/orlarge power variations. More specifically, since each switch operates atfixed duty cycle of 50 percent it may be easier to implement ZVStechniques. Other embodiments may use a different duty cycle algorithmto control the power delivered to the transmitter network and are withinthe scope of this disclosure.

Now referring to FIGS. 14B and 14C, other embodiments may employalternative circuits to achieve ZVS at light load conditions where theduty cycle may be small such that the switch pairs have minimal overlap.In such embodiments there may be inadequate energy stored in theresonant circuit inductor 260 to discharge the output capacitance (Coss)of switches 220, 225, 1205, 1210, so other circuit features may be addedto overcome this issue.

For example, circuit 1450 in FIGS. 14B, employs an alternativearrangement of resonant circuits that may improve ZVS at light loadconditions. Circuit 1450 is similar to circuit 1200 in FIG. 12, howevera second resonant circuit 1455 and a third resonant circuit 1460 havebeen added to each switching leg to generate additional inductivecurrent to discharge the output capacitance (Coss) of switches of eachcorresponding leg. In some embodiments resonant circuits 1455, 1460 maybe similar to resonant circuit 250, where their resonant frequency islower than the switching frequency. Regardless of the load or phasedelay of the switching pairs, each leg always switches at a 50 percentduty cycle, thus energizing second and third resonant circuits, 1455,1460, respectively, regardless of the duty cycle control. Operation ofthe switches and the ZVS timing is the same as employed in circuit 1200in FIG. 12, however circuit 1450 has additional load independent energyto ensure ZVS under all load conditions. Further, second and thirdresonant circuits, 1455, 1460, respectively, do not have an antennanetwork in series with them, so they may be designed with greaterflexibility than resonant circuit 250.

Now referring to circuit 1480 in FIG. 14C, another alternativeembodiment is illustrated that may improve ZVS at light load conditions.Circuit 1480 employs an additional resonant circuit 1485 in parallel tofirst resonant circuit 250. In some embodiments resonant circuit 1485may be similar to resonant circuit 250, where their resonant frequencyis lower than the switching frequency. In further embodiments thecurrent flowing through resonant circuit 1485 may be load or duty cycledependent, however it may still supply enough additional current toachieve ZVS. In further embodiments, operation of switches 220, 225,1205, 1210 and ZVS timing may be similar to circuit 1200 in FIG. 12,however circuit 1480 may have additional load independent energy toensure ZVS under all load conditions. Circuit 1480 may have fewercomponents than circuit 1450 in FIG. 14B. Resonant circuit 1485 may nothave an antenna network in series with it, so it may be designed withgreater flexibility than resonant circuit 250.

Now referring to FIG. 15, an example antenna coil 1500 of one embodimentis illustrated. Antenna coil 1500 may be used in an antenna network suchas, for example antenna network 405 in FIG. 12, to transmit electricalpower wirelessly to an electronic device. A receiver coil on the mobiledevice may be used to convert the wirelessly transferred power intoelectronic voltage and current. In one embodiment antenna coil 1500 isapproximately 65.5 mm long by 48.5 mm wide, however the size of theantenna coil may vary greatly in other embodiments.

Now referring to FIG. 16, a resonant half-bridge circuit 1600 that maybe used in some embodiments to convert a high voltage to a low voltagefor antenna network 405 is illustrated. Circuit 1600 is similar tocircuit 400 illustrated in FIG. 4, above, however in circuit 1600capacitor 410 (see FIG. 4) has been eliminated by carefully selectingresonant circuit 250 components inductor 1660 and capacitor 1665.Careful design and selection of inductor 1660 and capacitor 1665 mayresult in the appropriate AC voltage applied across antenna network 405or it may match the antenna network specification so that the antennanetwork can be directly connected across inductor 1660 or capacitor1665.

Now referring to FIG. 17, a resonant half-bridge circuit 1700 that maybe used in some embodiments to convert a high voltage to a low voltagefor antenna network 405 is illustrated. Circuit 1700 is similar tocircuit 500 illustrated in FIG. 5, above, however in circuit 1700inductor 505 (see FIG. 5) has been eliminated by carefully selectingresonant circuit 250 components inductor 1760 and capacitor 1765.Careful design and selection of inductor 1760 and capacitor 1765 mayresult in the appropriate AC voltage applied across antenna network 405or it may match the antenna network specification so that the antennanetwork can be directly connected across inductor 1760 or capacitor1765.

Now referring to FIG. 18, one embodiment of a receiver network 1800 isillustrated. Receiver network 1800 may be located in an electronicdevice and may receive transmitted electromagnetic energy from anantenna network. Receiver network 1800 may convert the receivedelectromagnetic energy into voltage and current that can be used tocharge and/or power the electronic device.

In one embodiment, receiver coil 1805 is exposed to transmittedelectromagnetic energy from a transmitter network as discussed above.Receiver coil 1805 coil may be used to generate an AC voltage withinnetwork 1800. In some embodiments, capacitor 1810 and receiver coil 1805form a resonant tank circuit that may be tuned to the transmitterfrequency to improve power transfer efficiency. In further embodiments,the coupled AC voltage may be converted to DC voltage through a fullwave bridge rectifier 1815 made of four diodes. The DC voltage can beconverted to other voltages to charge a battery or to power electroniccircuits.

Now referring to circuit 1900 in FIG. 19, in one embodiment a DC to DCconverter 1925 may be added to a receiver network to regulate its outputvoltage. In some embodiments, the coupled AC voltage through receivercoil 1905 may not be constant, and the voltage may change with factorssuch as magnetic field strength, distance between receiver andtransmitter coils, load and coil positions, among other factors.Therefore, the rectified DC voltage may vary and be essentiallyunregulated. For example, in one embodiment the output DC voltage mayvary from 5V to 25V. In some embodiments an unregulated DC output may betoo high for the load, thus a circuit such as DC to DC converter 1925may be used to regulate it.

Now referring to circuit 2000 illustrated in FIG. 20, some embodimentsmay improve the efficiency of a receiver network by replacing a diodebridge rectifier (e.g., rectifier 1815 in FIG. 18) with a synchronousswitch set 2025 comprising four solid-state switches. The use of switchset 2025 with a low on resistance may reduce losses due to thecomparatively larger voltage drop across the diodes in the diode bridge.In some embodiments the switches in switch set 2025 may turn on whencurrent flows through their body diodes from the source to the drain,and turn off when current flowing from their source to their drainreduces to zero or the current direction is reversed, flowing from theirdrain to their source. In further embodiments, synchronous switch set2025 can reduce two diode drops, (i.e., 1 V), which can improveefficiency by approximately 10% when the DC output is 10 V. However, insome embodiments a synchronous rectifier may still produce anunregulated DC output. If a regulated output voltage is required, suchembodiments may use a DC/DC converter such as converter 1925 illustratedin FIG. 19.

Now referring to FIGS. 21A and 21B, in some embodiments a receivernetwork may employ one or more bidirectional switches 2105, 2107, asdiscussed in more detail below. A bidirectional switch is a switch orswitch combination that enables the conduction of current in eitherdirection when in an on state and prevents the conduction of current inany direction when in an off state. In one embodiment, as illustrated inFIGS. 21A and 21B, a bidirectional switch 2105 may be made by connectinga first single switch 2110 and a second single switch 2115 back to backwith source terminals in common. Further embodiments may make abidirectional switch 2107 by connecting a first single switch 2110 and asecond single switch 2115 back to back with drain terminals in common,as illustrated in FIG. 21B. In some embodiments one or more of switches2105, 2107 may be silicon-based. In further embodiments, particularlywhen used in high frequency and/or high voltage applications, switches2105, 2107 may be GaN-based.

Now referring to FIG. 22, in some embodiments a receiver network circuit2200 may have two bidirectional switches 2210, 2215 combined with asynchronous rectifier 2220 to regulate DC output. In one embodiment,when the output voltage drops below a set low voltage threshold,switches 2210, 2215 and 2220 may operate like a normal synchronousrectifier. However, when the output voltage rises above a set highvoltage threshold, bidirectional switches 2210, 2215 may turn off, andblock the AC coil voltage from reaching the output capacitor.Bidirectional switches 2210, 2215 may remain off until the DC outputreduces below the set high voltage threshold, then the normalsynchronous mode may be reengaged. This embodiment may remove the needto have a separate DC/DC converter, and may improve efficiency and savecost.

Receiver network circuit 2300 illustrated in FIG. 23 has bidirectionalswitches 2310, 2315 on the bottom portion of receiver network circuit2300, as compared to circuit 2200 in FIG. 22 that has bidirectionalswitches 2210, 2215 on the top portion of the circuit. In furtherembodiments, as illustrated in FIG. 24, bidirectional switches 2410,2415 may be implemented on a first leg 2420 of receiver network circuit2400.

Now referring to FIG. 25, in further embodiments a receiver networkcircuit 2500 may employ a single stage regulation architecture toregulate output voltage. Such embodiments may not employ a synchronousrectifier and may replace two of the four diodes typically used in afull bridge rectifier with bidirectional switches 2410, 2415.Bidirectional switches 2410, 2515 may enable or disable the rectifier ondemand to control the output power and to regulate the DC voltage.Myriad other methods may be used for output power regulation and arewithin the scope of this disclosure.

Integration and Co-Packaging

Now referring to FIG. 26, in some embodiments one or more electroniccomponents may be integrated within a single electronic package 2600(i.e., co-packaged). In one embodiment a portion of half-bridge circuitsimilar to circuit 400 in FIG. 4 may be co-packaged by placing firstswitch 220 and second switch 225 in package 2600. In further embodimentsfirst switch 220 and second switch 225 may each have external source,gate and drain connections. An external connection may be an electricalconnection that is made outside of package 2600, such as a solderconnection to another circuit board. In other embodiments, first switch220 and second switch 225 may have external gate connections, howeverone or more of the source and drain connections may be inside package2600, forming a switch node connection. In some embodiments, the switchnode may also have an external connection.

In some embodiments electronic package 2600 may be what is known as anorganic multi-chip module. An organic substrate 2650, such as, but notlimited to a printed circuit board, may be used as a mount for theswitches 220, 225 and other components and may also provide electricalinterconnectivity between the devices within the package and/or betweenthe devices and the system to which package 2600 is mounted. In someembodiments one or more devices may be attached to the substrate with anelectrically conductive material such as, but not limited to, solder orelectrically conductive epoxy. In some embodiments the electronicdevices may be electrically connected to the substrate and/or each otherwith wire bonds, while in further embodiments flip-chip devices,conductive columns or other electrical interconnects may be used. Anelectrically insulative potting compound 2655 may be molded on top ofthe substrate and around the electrical devices to provide environmentalprotection.

In further embodiments, dies 220, 225 may each have a drivermonolithically integrated on the die. That is, first switch 220 may havea first driver circuit disposed on a unitary monolithic die. Similarly,second switch 225 may have a second driver circuit disposed on a unitarymonolithic die. In further embodiments, passive components such asresistors, capacitors, inductors and the like may also be mounted tosubstrate 2650. In yet further embodiments, additional active componentsmay be mounted to substrate such as diodes, a controller die or otherdevice.

In some embodiments, particularly in high frequency applications,co-packaging and monolithic integration may enable improved electricalperformance through the elimination of packaging and componentinterconnect parasitics. All conductors and electrical componentspossess parasitic elements. For instance, a resistor is designed topossess resistance, but will also possess unwanted parasiticcapacitance. Similarly, a conductor is designed to conduct an electricalsignal, but will also possess unwanted parasitic resistance andinductance. Parasitic elements cause propagation delays and impedancemismatches which limit the operating frequency of the converter. Thus,the elimination and or minimization of conductors and interconnectstructures between electronic components eliminates/minimizes parasiticelements that limit the maximum operating frequency of the converter.

Now referring to FIG. 27, first switch 220 and second switch 225 areshown in a different packaging configuration that may be called aleadless chip carrier or a quad flat no lead package 2700. First switch220 and second switch 225 are mounted to first pad 2710 and second pad2720, respectively, using methods as discussed above. Electricalconnections may be made from first switch 220 and second switch 225 tofirst pads 2710, between dies, to second pad 2720 and/or peripheralconnections 2725, as discussed above. First pad 2710, second pad 2720and peripheral connections 2727 may be leadframe material that is overmolded with an electrically insulative mold compound. First switch 220and second switch 225 may have monolithically integrated driver circuitsas discussed above. In further embodiments, other passive or activecomponents may also be integrated into package 2700, as discussed above.In one embodiment a full-bridge circuit may be made by mounting twopackages 2700 on a circuit board. In further embodiments, package 2600(see FIG. 26) or package 2700 may contain four separate switches suchthat all the switches necessary for a full-bridge converter are within asingle electronic package. In other embodiments a control die and otheractives may also be integrated within package 2700.

In the foregoing specification, embodiments of the invention have beendescribed with reference to numerous specific details that may vary fromimplementation to implementation. The specification and drawings are,accordingly, to be regarded in an illustrative rather than a restrictivesense. As an example, various embodiments may employ the same referencedesignation for circuit elements, however the circuit elementsthemselves may not be identical. One particular example is with regardto the antenna network. Both FIGS. 4 and 6 use the designation 405 forthe antenna network, however that does not imply the antenna network inFIG. 4 is identical to the antenna network in FIG. 6, it merelyillustrates that an antenna network is present in both embodiments. Thesole and exclusive indicator of the scope of the invention, and what isintended by the applicants to be the scope of the invention, is theliteral and equivalent scope of the set of claims that issue from thisapplication, in the specific form in which such claims issue, includingany subsequent correction. In addition, the scope of this disclosure isnot limited to a particular method of implementation. That is, thecircuits, the control algorithms and the packaging designs describedherein are not intended to be limiting, but are rather illustrativeexamples.

What is claimed is:
 1. A wireless power transmission circuit comprising:a voltage source having first and second output terminals; a firstsolid-state switch having a pair of first power terminals and a firstgate terminal, the pair of first power terminals connected between thefirst output terminal and a first switch node; a second solid-stateswitch having a pair of second power terminals and a second gateterminal, the pair of second power terminals connected between the firstswitch node and the second output terminal; a third solid-state switchhaving a pair of third power terminals and a third gate terminal, thepair of third power terminals connected between the first outputterminal and a second switch node; a fourth solid-state switch having apair of fourth power terminals and a fourth gate terminal, the pair offourth power terminals connected between the second switch node and thesecond output terminal; an antenna network coupled between the firstswitch node and the second switch node, and configured to transmitelectrical energy at a first frequency; and a controller coupled to thefirst, second, third and fourth gate terminals and configured to operatethe first, second, third and fourth solid-state switches such that theyeach operate at a fixed duty cycle and regulate power from the voltagesource to the antenna network by changing a phase delay between two ormore of the first, second, third and fourth solid-state switches.
 2. Thewireless power transmission circuit of claim 1 wherein in a firstportion of a switching cycle the first and second solid-state switchesoperate complementary and in a second portion of the switching cycle thethird and fourth solid-state switches operate complementary.
 3. Thewireless power transmission circuit of claim 2 wherein zero voltageswitching is performed in a deadtime located between the first portionof the switching cycle and the second portion of the switching cycle. 4.The wireless power transmission circuit of claim 1 wherein the powerfrom the voltage source is regulated by changing a phase delay betweenthe first and the third solid-state switches.
 5. The wireless powertransmission circuit of claim 1 wherein the first, second, third andfourth solid-state switches operate at a fixed duty cycle of 50 percent.6. The wireless power transmission circuit of claim 1 wherein the first,second, third and fourth solid-state switches are GaN-based devices. 7.The wireless power transmission circuit of claim 1 wherein the first,second, third and fourth solid-state switches operate at a switchingfrequency in a range of 1-20 MHz.
 8. The wireless power transmissioncircuit of claim 1 wherein the first, second, third and fourthsolid-state switches are configured to operate with a voltage potentialin the range of 50-1000 Volts DC.
 9. The wireless power transmissioncircuit of claim 1 further comprising a first resonant circuitconfigured to resonate at a second frequency and is coupled between thefirst switch node and the antenna network.
 10. The wireless powertransmission circuit of claim 9 further comprising an AC voltage dividercircuit coupled to the first resonant circuit and configured to reducean AC voltage supplied the antenna network.
 11. The wireless powertransmission circuit of claim 10 further comprising an impedancematching circuit coupled to the AC voltage divider circuit andconfigured to match an impedance of the antenna network.
 12. Thewireless power transmission circuit of claim 9 further comprising asecond resonant circuit coupled between the first switch node and thesecond output terminal; and a third resonant circuit coupled between thesecond switch node and the second output terminal.
 13. The wirelesspower transmission circuit of claim 9 further comprising a secondresonant circuit coupled in parallel with the first resonant circuit.14. A wireless power transmission circuit comprising: a voltage source;a full-bridge coupled to the voltage source and having first, second,third and fourth solid-state switches; an antenna network coupled to thefull-bridge rectifier and configured to transmit electrical energy at afirst frequency; and a controller coupled to the first, second, thirdand fourth solid-state switches and configured to regulate power fromthe voltage source to the antenna network by changing a phase delaybetween two or more of the first, second, third and fourth solid-stateswitches.
 15. The wireless power transmission circuit of claim 14wherein in a first portion of a switching cycle the first and secondsolid-state switches operate complementary and in a second portion of aswitching cycle the third and fourth solid-state switches operatecomplementary.
 16. The wireless power transmission circuit of claim 15wherein zero voltage switching is performed in a deadtime locatedbetween the first portion of the switching cycle and the second portionof the switching cycle.
 17. The wireless power transmission circuit ofclaim 14 wherein the first, second, third and fourth solid-stateswitches operate at a fixed duty cycle.
 18. The wireless powertransmission circuit of claim 17 wherein the fixed duty cycle isapproximately 50 percent.
 19. The wireless power transmission circuit ofclaim 14 wherein the phase delay is changed between the first and thethird solid-state switches.
 20. The wireless power transmission circuitof claim 14 wherein the first, second, third and fourth solid-stateswitches change state with zero voltage switching.
 21. A method ofoperating a wireless power transmission circuit, the method comprising:supplying power to the power transmission circuit with a voltage source;coupling power from the voltage source to the transmission circuit witha full-bridge comprising first, second, third and fourth solid-stateswitches; regulating power to the power transmission circuit with acontroller configured to change the power supplied to the powertransmission circuit by changing the phase delay between two or more ofthe first, second, third and fourth solid-state switches.
 22. The methodof claim 21 wherein in a first portion of a switching cycle the firstand second solid-state switches operate complementary and in a secondportion of the switching cycle the third and fourth solid-state switchesoperate complementary.
 23. The method of claim 22 wherein zero voltageswitching is performed in a deadtime located between the first portionof the switching cycle and the second portion of the switching cycle.24. The method of claim 21 wherein the first, second, third and fourthsolid-state switches operate at a fixed duty cycle.
 25. The method ofclaim 24 wherein the fixed duty cycle is 50 percent.
 26. The method ofclaim 21 wherein the phase delay is changed between the first and thethird solid-state switches.
 27. The method of claim 21 wherein thefirst, second, third and fourth solid-state switches change state withzero voltage switching.